Methods, systems and apparatus for controlling current supplied to control a machine

ABSTRACT

A current regulator is provided for an electric machine drive system for driving an electric machine. The current regulator includes an adjustable damping module that has a value of virtual damping resistance that is applied at the current regulator. The value of virtual damping resistance is adjustable as a function of sampling frequency. A controller can control the current regulator by determining whether the sampling frequency has changed since a previous execution cycle of the current regulator, and when the sampling frequency has changed since the previous execution cycle, the controller can modify the damping value as a function of the sampling frequency to allow the damping value to change with the sampling frequency. The damping value has a new value of virtual damping resistance that is applied at the current regulator after modifying the damping value. The controller can then execute the current regulator in accordance with the modified damping value to generate the voltage commands.

TECHNICAL FIELD

The present disclosure generally relates to techniques for controllingoperation of multi-phase systems that include alternating current (AC)machines, and more particularly relate to methods, systems and apparatusfor controlling current supplied to control an electric machine.

INTRODUCTION

Electric machines are utilized in a wide variety of applications. Forexample, hybrid/electric vehicles (HEVs) typically include an electrictraction drive system that includes a multi-phase alternating current(AC) electric motor which is driven by a power converter with a directcurrent (DC) power source, such as a storage battery. Motor windings ofthe AC electric motor can be coupled to inverter sub-modules of a powerinverter module (PIM). Each inverter sub-module includes a pair ofswitches that switch in a complementary manner to perform a rapidswitching function to convert the DC power to AC power. This AC powerdrives the AC electric motor, which in turn drives a shaft of HEV'sdrivetrain. For instance, some traditional HEVs implement twothree-phase pulse width modulated (PWM) inverter modules and twothree-phase AC machines (e.g., AC motors) each being driven by acorresponding one of the three-phase PWM inverter modules that it iscoupled to.

In such multi-phase systems, synchronous frame current regulators arecommonly used for current control of AC motors, such as three-phaseelectric motors. By providing dynamic control over a wide frequencyrange, synchronous frame current regulators are suited to manyindustrial applications. In digital implementations of conventionalcurrent regulators, as the ratio of the sampling frequency to thefundamental frequency, or synchronous frequency, of the AC motordecreases, the stability of these current regulators tends to decrease.For example, delays in digital implementation, increased sub-harmonicsin voltage synthesis using pulse width modulation (PWM), or the like,tend to introduce instability. To produce high torque within a limitedvolume, a high pole-count electric motor is useful, particularly forhybrid vehicle applications (e.g., hybrid electric vehicles or thelike). An increased pole-count generally increases the fundamentalfrequency associated with the AC motor, while the switching and samplingfrequency associated with the current regulation is generally limiteddue to limitations of the switching power device and the through-put ofthe processor. Typically, at maximum speed, the ratio of samplingfrequency to fundamental frequency, f_(samp)/f_(fund), can be very small(e.g., smaller than a ratio of about ten (10)). When this ratio is lessthan about ten (10), a discrete time domain controller may have asufficiently pronounced influence on the synchronous frame currentregulator. Furthermore, inner current loops associated with the currentregulator may incur instability due to digital delays. Sub-harmonicsassociated with asynchronous PWM become significant when the ratio islower than about twenty-one (21).

Some current regulators perform better under certain operatingconditions, while others perform better under other operatingconditions. Some current regulators implement a virtual dampingresistance to reduce parameter sensitivity and increase the disturbancerejection of the system, often described as increasing the stiffness ofthe drive system. In such current regulators, the virtual dampingresistance is set to a constant value. One drawback of this approach isthat the maximum achievable virtual damping resistance is limited to anextent by a minimum sampling frequency condition observed by the currentregulator.

Accordingly, it is desirable to provide methods and systems forcontrolling an AC motor that stabilize current regulation over a widerange of motor operating conditions. It would also be desirable toprovide current regulator architectures that can be modified duringdifferent operating conditions so that the current regulator topologybeing used performs well under the present operating condition.Additionally, it would be desirable to provide methods and systems forcurrent regulation of an AC motor that can operate as sampling frequencyvaries. Furthermore, other desirable features and characteristics of thepresent invention will become apparent from the subsequent detaileddescription and the appended claims, taken in conjunction with theaccompanying drawings and the foregoing technical field and background.

SUMMARY

In accordance with some of the disclosed embodiments, a currentregulator is provided for an electric machine drive system for drivingan electric machine. The current regulator includes an adjustabledamping module that has a value of virtual damping resistance that isapplied at the current regulator. The value of virtual dampingresistance is adjustable as a function of sampling frequency. Acontroller can control the current regulator by determining whether thesampling frequency has changed since a previous execution cycle of thecurrent regulator, and when the sampling frequency has changed since theprevious execution cycle, the controller can modify the damping value asa function of the sampling frequency to allow the damping value tochange with the sampling frequency. The damping value has a new value ofvirtual damping resistance that is applied at the current regulatorafter modifying the damping value. The controller can then execute thecurrent regulator in accordance with the modified damping value togenerate the voltage commands.

In one embodiment, the controller can store a previous damping valuethat is a previous value of the virtual damping resistance appliedduring the previous execution cycle of the current regulator, determinethe modified damping value based on the sampling frequency during acurrent execution cycle of the current regulator, and compute a changein damping value based on the difference between the modified dampingvalue and the previous damping value. The modified damping value is anew value of virtual damping resistance based on a new sample frequency,and the change in the damping value is a change in the virtual dampingresistance based on a difference between the new value of virtualdamping resistance and the previous value of virtual damping resistance.

In one embodiment, the current regulator includes integrators, and thecontroller can reinitialize, prior to executing the current regulator,integration terms generated by the integrators of the current regulatorwhen the virtual damping resistance is updated. In one embodiment, thecurrent regulator can generate current error values and apply gains tothe current error values prior to providing the current error values tothe integrators that generate integration terms. Each current errorvalue is determined based on a difference between a current commandvalue and a stator current value from the electric machine. In oneembodiment, the current regulator can generate voltage commands for thecurrent execution cycle based on the modified damping value and updatedvalues of integration terms generated by the integrators by setting eachto presently determined values.

In one embodiment, the electric machine and the electric machine drivesystem including the current regulator can be implemented within avehicle.

In one embodiment, the controller can perform a method to control thecurrent regulator. In accordance with the method, the controller candetermine whether a sampling frequency has changed since a previousexecution cycle of the current regulator. When the sampling frequencyhas changed since the previous execution cycle, the controller canmodify a damping value of the current regulator as a function ofsampling frequency to allow the damping value to change with thesampling frequency such that the damping value has a new value ofvirtual damping resistance that is applied at the current regulatorafter modifying. The controller can then execute the current regulatorto generate the voltage commands in accordance with the modified dampingvalue. The controller can modify the damping value of the currentregulator by updating and setting the damping value as a function of thesampling frequency. For instance, in one embodiment, the controller cancompute the new value of virtual damping resistance using an equation,and in another embodiment, the controller can determine the new value ofvirtual damping resistance via a lookup table.

In one embodiment, the controller can store a previous damping valuethat is a previous value of the virtual damping resistance appliedduring the previous execution cycle of the current regulator; determinethe modified damping value based on the sampling frequency during acurrent execution cycle of the current regulator; and compute a changein damping value based on the difference between the modified dampingvalue and the previous damping value. The modified damping value is anew value of virtual damping resistance based on a new sample frequency,and the change in the damping value is a change in the virtual dampingresistance based on a difference between the new value of virtualdamping resistance and the previous value of virtual damping resistance.

In one embodiment, the current regulator can generate current errorvalues, and apply gains to the current error values prior to providingthe current error values to integrators of the current regulator. Eachcurrent error value is determined based on a difference between acurrent command value and a stator current value from the electricmachine. In one embodiment, prior to executing the current regulatorduring a current execution cycle of the current regulator, thecontroller can reinitialize integration terms generated by integratorsof the current regulator when the virtual damping resistance is updated.For example, the controller can reinitialize integration terms generatedby integrators during a transition period to avoid inducing disturbancevoltages as when a previous damping value is being updated to themodified damping value. In one embodiment, the controller can executethe current regulator using the modified damping value and updatedvalues of the integration terms generated by the integrators by settingeach to presently determined values to generate voltage commands for thecurrent execution cycle of the current regulator.

BRIEF DESCRIPTION OF THE DRAWINGS

The exemplary embodiments will hereinafter be described in conjunctionwith the following drawing figures, wherein like numerals denote likeelements, and wherein:

FIG. 1 illustrates one non-limiting example, of a vehicle in which thedisclosed embodiments may be implemented.

FIG. 2 is a block diagram of one example of a vector controlled motordrive system in accordance with various embodiments.

FIG. 3 is a block diagram of a portion of a motor drive system includinga three-phase voltage source inverter module connected to a three-phaseAC motor.

FIGS. 4A and 4B are block diagrams of a current regulator in accordancewith one implementation of the disclosed embodiments.

FIG. 5 is a flowchart that illustrates a control method that can beapplied to at a current regulator of FIGS. 4A and 4B in accordance withthe disclosed embodiments.

FIG. 6 is a graph that illustrates how the current regulator can beconfigured to operate in a first operational mode in FIG. 4A when thespeed is greater than a first speed threshold (ω₂), or can be configuredto operate in a second operational mode shown in FIG. 4B when the speedis below a second speed threshold (ω₁) in accordance with the disclosedembodiments.

FIG. 7 is a block diagram of a current regulator in accordance withanother implementation of the disclosed embodiments.

FIG. 8 is a flowchart that illustrates another control method that canbe applied to at a current regulator of FIG. 7 in accordance with thedisclosed embodiments.

FIG. 9 is a graph that illustrates how virtual damping resistance(Rdamp) varies as a function sampling frequency (F_(S)) in accordancewith the embodiment illustrated in FIGS. 7 and 8.

DETAILED DESCRIPTION

The following detailed description is merely exemplary in nature and isnot intended to limit the application and uses. Furthermore, there is nointention to be bound by any expressed or implied theory presented inthe preceding technical field, background, brief summary or thefollowing detailed description. As used herein, the term module refersto any hardware, software, firmware, electronic control component,processing logic, and/or processor device, individually or in anycombination, including without limitation: application specificintegrated circuit (ASIC), an electronic circuit, a processor (shared,dedicated, or group) and memory that executes one or more software orfirmware programs, a combinational logic circuit, and/or other suitablecomponents that provide the described functionality.

Embodiments of the present disclosure may be described herein in termsof functional and/or logical block components and various processingsteps. It should be appreciated that such block components may berealized by any number of hardware, software, and/or firmware componentsconfigured to perform the specified functions. For example, anembodiment of the present disclosure may employ various integratedcircuit components, e.g., memory elements, digital signal processingelements, logic elements, look-up tables, or the like, which may carryout a variety of functions under the control of one or moremicroprocessors or other control devices. In addition, those skilled inthe art will appreciate that embodiments of the present disclosure maybe practiced in conjunction with any number of systems, and that thesystems described herein are merely exemplary embodiments of the presentdisclosure.

For the sake of brevity, conventional techniques related to signalprocessing, data transmission, signaling, control, and other functionalaspects of the systems (and the individual operating components of thesystems) may not be described in detail herein. Furthermore, theconnecting lines shown in the various figures contained herein areintended to represent example functional relationships and/or physicalcouplings between the various elements. It should be noted that manyalternative or additional functional relationships or physicalconnections may be present in an embodiment of the present disclosure.

As used herein, the word “exemplary” means “serving as an example,instance, or illustration.” The following detailed description is merelyexemplary in nature and is not intended to limit the invention or theapplication and uses of the invention. Any embodiment described hereinas “exemplary” is not necessarily to be construed as preferred oradvantageous over other embodiments. All of the embodiments described inthis Detailed Description are exemplary embodiments provided to enablepersons skilled in the art to make or use the invention and not to limitthe scope of the invention which is defined by the claims. Furthermore,there is no intention to be bound by any expressed or implied theorypresented in the preceding technical field, background, brief summary orthe following detailed description.

Before describing in detail embodiments that are in accordance with thepresent invention, it should be observed that the embodiments resideprimarily in combinations of method steps and apparatus componentsrelated to controlling operation of a multi-phase system. It will beappreciated that embodiments of the invention described herein can beimplemented using hardware, software or a combination thereof. Thecontrol circuits described herein may comprise various components,modules, circuits and other logic which can be implemented using acombination of analog and/or digital circuits, discrete or integratedanalog or digital electronic circuits or combinations thereof. As usedherein the term “module” refers to a device, a circuit, an electricalcomponent, and/or a software based component for performing a task. Insome implementations, the control circuits described herein can beimplemented using one or more application specific integrated circuits(ASICs), one or more microprocessors, and/or one or more digital signalprocessor (DSP) based circuits when implementing part or all of thecontrol logic in such circuits. It will be appreciated that embodimentsof the invention described herein may be comprised of one or moreconventional processors and unique stored program instructions thatcontrol the one or more processors to implement, in conjunction withcertain non-processor circuits, some, most, or all of the functions forcontrolling operation of a multi-phase system, as described herein. Assuch, these functions may be interpreted as steps of a method forcontrolling operation of a multi-phase system. Alternatively, some orall functions could be implemented by a state machine that has no storedprogram instructions, or in one or more application specific integratedcircuits (ASICs), in which each function or some combinations of certainof the functions are implemented as custom logic. Of course, acombination of the two approaches could be used. Thus, methods and meansfor these functions will be described herein. Further, it is expectedthat one of ordinary skill, notwithstanding possibly significant effortand many design choices motivated by, for example, available time,current technology, and economic considerations, when guided by theconcepts and principles disclosed herein will be readily capable ofgenerating such software instructions and programs and ICs with minimalexperimentation.

Overview

Some current regulators perform better under certain operatingconditions, while others perform better under other operatingconditions. For example, a state feedback decoupling (SFbD) performswell under high acceleration conditions (e.g., vehicle shudder,driveline-resonance, wheel slip conditions) and when considering theeffects of discrete control. By contrast, a complex vector currentregulator (CVCR) performs well when operating under low and moderateacceleration conditions, but over a very wide speed range, includinghigh speeds with overmodulation and six-step control operatingconditions. The disclosed embodiments provide a current regulatorarchitecture that can be varied to allow for multiple different currentregulator topologies to be used based on operating conditions where aparticular current regulator topology performs best. The disclosedembodiments can allow for the appropriate current regulatorconfiguration to be selected based on speed (e.g., motor speed orelectrical fundamental frequency), and then instantaneously, andsmoothly, transition between the state feedback decoupling (SFbD)current regulator configuration and the complex vector current regulatorconfiguration at set speed breakpoints. The disclosed embodiments canutilize each current regulator configuration where it is mostadvantageous to improve current control during high-accelerationconditions without degrading performance in the overmodulation andsix-step operating regions.

In one embodiment, a current regulator is provided for an electricmachine drive system for driving an electric machine. The currentregulator is configurable to operate in a first configuration or asecond configuration depending on a synchronous speed of the electricmachine (or other machine or drive states). A controller can configurean operational mode of the current regulator by selecting, based on thesynchronous speed of the electric machine, either the firstconfiguration of the current regulator or the second configuration ofthe current regulator as a currently active configuration, and can thenexecute the current regulator in accordance with the currently activeconfiguration. The first configuration of the current regulatorcomprises a first set of elements and cross-coupling gain blocks,whereas the second configuration of the current regulator can includethe first set of elements without the cross-coupling gain blocks. Thefirst set elements can vary depending on the implementation, but cangenerally include: summing junctions, integrators, and gain blocks. Inone embodiment, the current regulator is configured to operate as acomplex vector current regulator when configured in the firstconfiguration, and is configured to operate as a state feedbackdecoupling (SFbD) current regulator when configured in the secondconfiguration.

Embodiments of the present invention relate to methods, systems andapparatus for controlling operation of a multi-phase system that can beimplemented, for example, in operating environments such as ahybrid/electric vehicle (HEV). Embodiments of the present inventionrelate to methods, systems and apparatus for current regulators. In theexemplary implementations which will now be described, the controltechniques and technologies will be described as applied to ahybrid/electric vehicle. However, it will be appreciated by thoseskilled in the art that the same or similar techniques and technologiescan be applied in the context of other systems in which it is desirableto control operation of a multi-phase system. In this regard, any of theconcepts disclosed here can be applied generally to “vehicles,” and asused herein, the term “vehicle” broadly refers to a non-living transportmechanism having an AC machine. In addition, the term “vehicle” is notlimited by any specific propulsion technology such as gasoline or dieselfuel. Rather, vehicles also include hybrid vehicles, battery electricvehicles, hydrogen vehicles, and vehicles which operate using variousother alternative fuels.

As used herein, the term “alternating current (AC) machine” generallyrefers to “a device or apparatus that converts electrical energy tomechanical energy or vice versa.” AC machines can generally beclassified into synchronous AC machines and asynchronous AC machines.Synchronous AC machines can include permanent magnet machines andreluctance machines. Permanent magnet machines include surface mountpermanent magnet machines (SMPMMs) and interior permanent magnetmachines (IPMMs). Although an AC machine can be an AC motor (e.g.,apparatus used to convert AC electrical energy or power at its input toproduce to mechanical energy or power), an AC machine is not limited tobeing an AC motor, but can also encompass generators that are used toconvert mechanical energy or power at its prime mover into electrical ACenergy or power at its output. Any of the machines can be an AC motor oran AC generator. An AC motor is an electric motor that is driven by analternating current. In some implementations, an AC motor includes anoutside stationary stator having coils supplied with alternating currentto produce a rotating magnetic field, and an inside rotor attached tothe output shaft that is given a torque by the rotating field.

FIG. 1 illustrates one non-limiting example, of a vehicle, or automobile1, in which the disclosed embodiments may be implemented. The automobile1 includes a driveshaft 12, a body 14, four wheels 16, and an electroniccontrol system 18. The body 14 is arranged on the chassis andsubstantially encloses the other components of the automobile 1. Thebody 14 and the chassis may jointly form a frame. The wheels 16 are eachrotationally coupled to the chassis near a respective corner of the body14.

The automobile 1 may be any one of a number of different types ofautomobiles, such as, for example, a sedan, a wagon, a truck, or a sportutility vehicle (SUV), and may be two-wheel drive (2WD) (i.e.,rear-wheel drive or front-wheel drive), four-wheel drive (4WD), orall-wheel drive (AWD). The automobile 1 may also incorporate any one of,or combination of, a number of different types of engines, such as, forexample, a gasoline or diesel fueled combustion engine, a “flex fuelvehicle” (FFV) engine (i.e., using a mixture of gasoline and alcohol), agaseous compound (e.g., hydrogen and natural gas) fueled engine, acombustion/electric motor hybrid engine, and an electric motor.

In the exemplary embodiment illustrated in FIG. 1, the automobile 1further includes a motor 20 (i.e., an electric motor/generator, tractionmotor, etc.), energy sources 22, 24, and a power inverter assembly 10.As shown in FIG. 1, the motor 20 may also include a transmissionintegrated therein such that the motor 20 and the transmission aremechanically coupled to at least some of the wheels 16 through one ormore half shafts 30.

As shown, the energy source 22, 24 are in operable communication and/orelectrically coupled to the electronic control system 18 and the powerinverter assembly 10. Although not illustrated, the energy sources 22,24 may vary depending on the embodiment and may be of the same ordifferent type. In one or more embodiments, the energy sources 22, 24may each comprise a battery, a fuel cell, an ultracapacitor, or anothersuitable voltage source. A battery may be any type of battery suitablefor use in a desired application, such as a lead acid battery, alithium-ion battery, a nickel-metal battery, or another rechargeablebattery. An ultracapacitor may comprise a supercapacitor, anelectrochemical double layer capacitor, or any other electrochemicalcapacitor with high energy density suitable for a desired application.

The motor 20 can be a multi-phase alternating current (AC) motor andincludes windings, where each winding corresponds to one phase of themotor 20, as will be described in greater detail below. Although notillustrated, the motor 20 can include a stator assembly (including thecoils), a rotor assembly (e.g., including a ferromagnetic core), and acooling fluid (i.e., coolant), as will be appreciated by one skilled inthe art. The motor 20 may be an induction motor, a permanent magnetmotor, a synchronous reluctance motor, or any type suitable for thedesired application.

FIG. 2 is a block diagram of one example of a vector controlled motordrive system 100 in accordance with the disclosed embodiments. Thesystem 100 controls a three-phase AC machine 120 via a three-phase pulsewidth modulated (PWM) inverter module 110 coupled to the three-phase ACmachine 120 so that the three-phase AC machine 120 can efficiently use aDC input voltage (Vdc) provided to the three-phase PWM inverter module110 by adjusting current commands that control the three-phase ACmachine 120. The three-phase AC machine 120 can be used as the motor 20of FIG. 1. In one particular implementation, the vector controlled motordrive system 100 can be used to control torque in an HEV.

In the following description of one particular non-limitingimplementation, the three-phase AC machine 120 is described as athree-phase AC powered motor 120, and in particular a three-phase,permanent magnet synchronous AC powered motor (or more broadly as amotor 120); however, it should be appreciated that the illustratedembodiment is only one non-limiting example of the types of AC machinesthat the disclosed embodiments can be applied to, and further that thedisclosed embodiments can be applied to any type of multi-phase ACmachine that includes fewer or more phases.

The three-phase AC motor 120 is coupled to the three-phase PWM invertermodule 110 via three inverter poles and generates mechanical power(Torque X Speed) based on three-phase sinusoidal current signalsreceived from the PWM inverter module 110. In some implementations, theangular position of a rotor (θr) of the three-phase AC motor 120 or“shaft position” is measured using a position sensor (not illustrated),and in other implementations, the angular position of a rotor (θr) ofthe three-phase AC motor 120 can be estimated without using a positionsensor by using sensorless position estimation techniques.

Prior to describing operation details of the system 100, a more detaileddescription of one exemplary implementation of the three-phase voltagesource inverter 110 will be provided (including how it is connected tothe three-phase AC motor 120) with reference to FIG. 3.

FIG. 3 is a block diagram of a portion of a motor drive system includinga three-phase voltage source inverter 110 connected to a three-phase ACmotor 120. It should be noted that the three-phase voltage sourceinverter 110 and the three-phase motor 120 in FIG. 2 are not limited tothis implementation; rather, FIG. 3 is merely one example of how thethree-phase voltage source inverter 110 and the three-phase motor 120 inFIG. 2 could be implemented in one particular embodiment.

As illustrated in FIG. 3, the three-phase AC motor 120 has three statoror motor windings 120 a, 120 b, 120 c, connected to motor terminals A,B, C, and the three-phase PWM inverter module 110 includes a capacitor270 and three inverter sub-modules 115-117. In this particularembodiment, in phase A the inverter sub-module 115 is coupled to motorwinding 120 a, in phase B the inverter sub-module 116 is coupled tomotor winding 120 b, and in phase C the inverter sub-module 117 iscoupled to motor winding 120 c. The current flow is bi-directional intoand out of all motor windings 120.

The resultant phase or stator currents (Ias-Ics) 122, 123, 124, flowthrough respective stator windings 120 a-c. The phase to neutralvoltages across each of the stator windings 120 a-120 c are respectivelydesignated as VAN, VBN, VCN, with the back electromotive force (EMF)voltages generated in each of the stator windings 120 a-120 crespectively shown as the voltages E_(a), E_(b), E_(c), produced byideal voltage sources, each respectively shown connected in series withstator windings 120 a-120 c. As is well known, these back-EMF voltagesE_(a), E_(b), E_(c), are the voltages induced in the respective statorwindings 120 a-120 c by the rotation of the rotor magnetic field.Although not shown, the motor 120 is coupled to a drive shaft.

The inverter 110 includes a capacitor 270, a first inverter sub-module115 comprising a dual switch 272/273, 274/275, a second invertersub-module 116 comprising a dual switch 276/277, 278/279, and a thirdinverter sub-module 117 comprising a dual switch 280/281, 282/283. Assuch, inverter 110 has six solid state controllable switching devices272, 274, 276, 278, 280, 282, and six diodes 273, 275, 277, 279, 281,283, to appropriately switch DC voltage (V_(DC)) and provide three-phaseenergization of the stator windings 120 a, 120 b, 120 c of thethree-phase AC motor 120.

A high-level motor controller 112 can receive motor command signals andmotor operating signals from the motor 120, and generate control signalsfor controlling the switching of solid state switching devices 272, 274,276, 278, 280, 282 within the inverter sub-modules 115-117. By providingappropriate control signals to the individual inverter sub-modules115-117, the controller 112 controls switching of solid state switchingdevices 272, 274, 276, 278, 280, 282, within the inverter sub-modules115-117 and thereby controls the outputs of the inverter sub-modules115-117 that are provided to motor windings 120 a-120 c, respectively.The resultant stator currents (Ias . . . Ics) 122-124 that are generatedby the inverter sub-modules 115-117 of the three-phase inverter module110 are provided to motor windings 120 a, 120 b, 120 c. The voltages asV_(AN), V_(BN), V_(CN), and the voltage at node N fluctuate over timedepending on the open/close states of switches 272, 274, 276, 278, 280,282 in the inverter sub-modules 115-117 of the inverter module 110, aswill be described below.

Referring again to FIG. 2, the vector control motor drive system 100includes a torque-to-current mapping module 140, a synchronous (SYNC.)frame current regulator module 170, a synchronous-to-stationary(SYNC.-TO-STAT.) transformation module 102, an αβ reference frame-to-abcreference frame (αβ-to-abc) transformation module 106, a pulse widthmodulation (PWM) module 108, a three-phase PWM inverter 110, an abcreference frame-to-αβ reference frame (abc-to-αβ) transformation module127, and a stationary-to-synchronous (STAT.-TO-SYNC.) transformationmodule 130.

The torque-to-current mapping module 140 receives a torque commandsignal (Te*) 136, angular rotation speed (ωr) 138 of the shaft that isgenerated based on the derivative of the rotor/shaft position output(θr) 121, and the DC input voltage (V_(DC)) 139 as inputs, along withpossibly a variety of other system parameters depending uponimplementation. The torque-to-current mapping module 140 uses theseinputs to generate a d-axis current command (Id*) 142 and a q-axiscurrent command (Iq*) 144 that will cause the motor 120 to generate thecommanded torque (Te*) at motor speed (ωr) 138. In particular, thetorque-to-current mapping module 140 uses the inputs to map the torquecommand signal (Te*) 136 to a d-axis current command signal (Id*) 142and a q-axis current command signal (Iq*) 144. The synchronous referenceframe d-axis and q-axis current command signals (Id*, Iq*) 142, 144 areDC commands that have a constant value as a function of time duringsteady-state operation.

Blocks 127, 130 collectively make up a reverse transformation modulethat can transform AC signals (e.g., the three-phase sinusoidal statorcurrents) into DC Cartesian signals (e.g., a d-axis synchronous framestator current and a q-axis synchronous frame stator current) for use bythe current regulator 170. In one embodiment, a detector (not shown) maybe coupled to the AC motor 120 to sample the AC signals and supply theseand other measured quantities (e.g., from a variety of system outputs)to the current regulator 170 (and other high-level controllers). Forexample, the detector may measure a supply potential (e.g., a batterypotential or DC bus voltage (V_(dc))), the phase or stator currents, amotor speed (ω_(r)) 138 of the AC motor 120, a rotor phase angle (θr)121 of the AC motor 120, or the like.

In one embodiment, the abc-to-αβ transformation module 127 receives themeasured three-phase stationary reference frame feedback stator currents(Ias . . . Ics) 122-124 that are fedback from motor 120. The abc-to-αβtransformation module 127 uses these three-phase stationary referenceframe feedback stator currents 122-124 to perform an abc referenceframe-to-αβ reference frame transformation to transform the three-phasestationary reference frame feedback stator currents 122-124 intostationary reference frame feedback stator currents (Iα, Iβ) 128, 129.The abc-to-αβ transformation is well-known in the art and for sake ofbrevity will not be described in detail. The stationary-to-synchronoustransformation module 130 receives the stationary reference framefeedback stator currents (Iα, Iβ) 128, 129 and the rotor angularposition (θr) 121 and generates (e.g., processes or converts) thesestationary reference frame feedback stator currents (Iα, Iβ) 128, 129 togenerate a synchronous reference frame d-axis current signal (Id) 132and a synchronous reference frame q-axis current signal (Iq) 134. Theprocess of stationary-to-synchronous conversion is well-known in the artand for sake of brevity will not be described in detail.

The high-level controller 112 executes one or more programs (e.g., tooptimize commanded currents for a predetermined control parameter, orthe like) to determine operating inputs (e.g., modified commandedcurrents, commanded voltages, torque commands, or the like) used forcontrolling the AC motor 120 via the current regulator 170. In addition,the high-level controller can execute logic to control the currentregulator 170 as will be described below in detailed with reference toFIGS. 4A through 9.

One or more of the components of the controller 112 may be embodied insoftware or firmware, hardware, such as an application specificintegrated circuit (ASIC), an electronic circuit, a processor (shared,dedicated, or group) and memory that execute one or more software orfirmware programs, a combinational logic circuit, and/or other suitablecomponents, or a combination thereof. In one embodiment, the controller112 is partitioned into one or more processing modules that areassociated with one or more of the controller operations. For example,the current regulator 170 may be implemented as one of these processingmodules. Although not shown, the controller 112 may include additionalmodules, such as a commanded current source, a torque module, afield-weakening voltage control module, an overmodulation module or thelike. Additionally, one or more of the various processing modules of thecontroller 112, as well as one or more of the operations of thecontroller 112, may be embodied as separate components of the drivesystem 100 or incorporated with another component of the drive system100.

Generally, the current regulator 170 produces commanded voltages andsupplies the commanded voltages to the inverter 110 through blocks 102,106, 108, which collectively make up a transformation module. Thecurrent regulator 170 produces direct current (DC) Cartesian commandedvoltages (e.g., a d-axis synchronous frame commanded voltage and aq-axis synchronous frame commanded voltage) in steady-state. Thetransformation module 102, 106, 108 converts the DC Cartesian commandedvoltages to three-phase AC commanded voltages (e.g., a first phasecommanded voltage (v_(as)*), a second phase commanded voltage (v_(bs)*),and a third phase commanded voltage (v_(cs)*)) and supplies thethree-phase AC commanded voltages to the inverter 110.

In one embodiment, to produce the d- and q-axis commanded voltages 172,174, the current regulator 170 utilizes several inputs. For example, thecurrent regulator 170 can use current signals 132, 134 (d-axis andq-axis synchronous frame stator currents), the commanded currents 142,144, and decoupling voltages (not shown in FIG. 2) provided by thecontroller 112 to produce the d- and q-axis commanded voltages 172, 174.For example, the controller 112 may retrieve the commanded currents froma commanded current table 140 that can be stored in a memory of thecontroller 112. The commanded current table is preferably optimized forone or more pre-determined control parameters (e.g., system efficiency)and may be derived from any number of models for optimizing the desiredcontrol parameter(s). Additionally, the commanded current table may bepre-determined based on voltage and current limits of the AC motor 120such that the commanded current source applies an appropriate amount ofd-axis and q-axis currents to the AC motor 120 to produce a desiredtorque (e.g., with high efficiency) and maintain current regulationstability. The inverter voltage limits may be pre-determined based onthe supply voltage or calculated online, and the feedforward terms maybe determined by the controller 112 based on the d-axis and q-axissynchronous frame stator currents, the motor speed, and the motorparameters.

In this embodiment, the synchronous frame current regulator module 170receives the synchronous reference frame d-axis current signal (Id) 132,the synchronous reference frame q-axis current signal (Iq) 134, thed-axis current command (Id*) 142 and the q-axis current command (Iq*)144, and uses these signals to generate a synchronous reference framed-axis voltage command signal (Vd*) 172 and a synchronous referenceframe q-axis voltage command signal (Vq*) 174. The synchronous referenceframe voltage command signals (Vd*, Vq*) 172, 174 are commands that havea constant value as a function of time when in steady state operation.The synchronous frame current regulator module 170 outputs thesynchronous reference frame d-axis voltage command signal (Vd*) 172 andthe synchronous reference frame q-axis voltage command signal (Vq*) 174.Further detail regarding operation of the synchronous frame currentregulator module 170 and the process of current to voltage conversionwill be described in greater detail below with reference to FIGS. 4Athrough 9. Because the current commands are DC signals in thesynchronous reference frame during steady-state they are easier toregulate in comparison to AC stationary reference frame currentcommands.

The synchronous-to-stationary transformation module 102 receives thevoltage command signals (Vd *, Vq *) 172, 174 as inputs along with therotor position output (θr) 121. In response to the voltage commandsignals (Vd*, Vq*) 172, 174 and the measured (or estimated) rotorposition angle (θr) 121, the synchronous-to-stationary transformationmodule 102 performs a dq-to-αβ transformation to generate an α-axisstationary reference frame voltage command signal (Vα*) 104 and a β-axisstationary reference frame voltage command signal (Vβ*) 105. Thestationary reference frame α-axis and β-axis voltage command signals(Vα*, Vβ*) 104, 105 are in the stationary reference frame and thereforehave values that vary as a sine wave as a function of time in steadystate. The process of synchronous-to-stationary conversion is well-knownin the art and for sake of brevity will not be described in detail.

The αβ-to-abc transformation module 106 receives the stationaryreference frame voltage command signals (Vα*, Vβ*) 104, 105, and basedon these signals, generates stationary reference frame voltage commandsignals (Vas* . . . Vcs*) 107 (also referred to as “phase voltagecommand signals”) that are sent to the PWM module 108. The αβ-to-abctransformation is well-known in the art and for sake of brevity will notbe described in detail.

The three-phase PWM inverter module 110 is coupled to the PWM module108. The PWM module 108 is used for the control of pulse widthmodulation (PWM) of the phase voltage command signals (Vas* . . . Vcs*)107. The switching vector signals (Sa . . . Sc) 109 are generated basedon duty cycle waveforms that are not illustrated in FIG. 2, but areinstead internally generated at the PWM module 108 to have a particularduty cycle during each PWM period. The PWM module 108 utilizes the phasevoltage command signals (Vas* . . . Vcs*) 107 to calculate the dutycycle waveforms (not illustrated in FIG. 2) to generate switching vectorsignals (Sa . . . Sc) 109, which it provides to the three-phase PWMinverter module 110. The particular modulation algorithm implemented inthe PWM module 108 can be any known modulation algorithm including SpaceVector Pulse Width Modulation (SVPWM) techniques to control of pulsewidth modulation (PWM) to create alternating current (AC) waveforms thatdrive the three-phase AC powered machine 120 based on the DC input 139.

The switching vector signals (Sa . . . Sc) 109 control the switchingstates of switches in PWM inverter 110 to generate the commandedthree-phase voltage at each phase A, B, C. The switching vector signals(Sa . . . Sc) 109 are PWM waveforms that have a particular duty cycleduring each PWM period that is determined by the duty cycle waveformsthat are internally generated at the PWM module 108.

The inverter 110 (e.g., a pulse width modulation (PWM) voltage sourceinverter (VSI)) is coupled to the machine 120. In response to thecommanded voltages and a supply potential (V_(dc)), the inverter 110produces AC voltages which are used to drive the AC motor 120. As aresult, stator currents are generated in the windings of the AC motor120. The inverter 110 can also vary the amount of AC voltage applied tothe AC motor 120 (e.g., the inverter 110 can vary the voltage usingPWM), thus allowing the controller 12 to control the AC motor current.For example, the amount of voltage that the inverter 110 applies to theAC motor 120 may be indicated by a modulation index, and the PWM may beestablished between pre-determined modulation index limits. In oneembodiment, synchronous PWM is utilized to vary the amount of AC voltageapplied to the AC motor 120, although other PWM techniques may also beused.

The three-phase PWM inverter module 110 receives the DC input voltage(Vdc) and switching vector signals (Sa . . . Sc) 109, and uses them togenerate three-phase alternating current (AC) voltage signal waveformsat inverter poles that drive the three-phase AC machine 120 at varyingspeeds (ωr). The three-phase machine 120 receives the three-phasevoltage signals generated by the PWM inverter 110 and generates a motoroutput at the commanded torque Te* 136. In this particularimplementation, the machine 120 comprises a three-phase interiorpermanent-magnet synchronous motor (IPMSM) 120, but the disclosedembodiments can be any multi-phase AC machine having any number ofphases.

Although not illustrated in FIG. 2, the system 100 may also include agear coupled to and driven by a shaft of the three-phase AC machine 120.The measured feedback stator currents (Ia-Ic) 122-124 are sensed,sampled and provided to the abc-to-αβ transformation module 127 asdescribed above.

Current Regulator Module

In accordance with the disclosed embodiments, methods are provided forinstantaneously and smoothly transitioning between current regulatortopologies as a function of motor speed in this example, but could bebased on other equivalent motor states, such as electrical fundamentalfrequency, inverter switching frequency or motor acceleration. Forexample, in one embodiment, a method is provided for instantaneously andsmoothly transitioning between a State Feedback Decoupling (SFbD)current regulator and a Complex Vector Current Regulator (CVCR) as afunction of motor speed. The method utilizes each type of currentregulator during the operating conditions where it is most advantageousand where it performs best. For example, SFbD current regulator performswell under high acceleration conditions (e.g., vehicle shudder,driveline-resonance, wheel slip conditions) and when considering theeffects of discrete control. The CVCR performs well in loweracceleration conditions, and over a wider range of speed including theovermodulation and six-step control regions. The method can select,based on motor speed, which type of current regulator should be activeand transition between the different types of current regulators (e.g.,at set speed breakpoints). The disclosed embodiments can allow eachcurrent regulator topology to be used in the operating space where itperforms best to improve current control during high-accelerationconditions without degrading performance in the overmodulation andsix-step operating regions. Improved current control can improve vehicledrive, as exhibited, for example, by noise, vibration or harshness (NVH)(e.g., eliminate or reduce vehicle low-speed shudder) when using theSFbD current regulator, and there is no negative impact onovermodulation and six-step control by maintaining use of CVCR at higherspeeds.

FIGS. 4A and 4B are block diagrams of a current regulator module 170 inaccordance with one implementation of the disclosed embodiments. Thecurrent regulator module 170 includes various blocks shown in FIG. 4A.When all blocks are enabled, the current regulator module 170 functionsas a complex vector (CV) current regulator module 170-1. By contrast,when certain blocks 324, 344 are disabled (e.g., when gains Kppd, Kppqare set to zero), as shown in FIG. 4B, the current regulator module 170functions as a state feedback decoupling (SFbD) current regulator module170-2. As will be described in greater detail below, in accordance withthe disclosed embodiments, techniques are provided for switchingoperational modes of the current regulator module 170, based on speed(e.g., motor speed or electrical fundamental frequency) breakpoints, tochange its topology (in terms of active blocks) so that it functions aseither a complex vector (CV) current regulator module 170-1 or a statefeedback decoupling (SFbD) current regulator module 170-2. In otherwords, based on speed breakpoints, certain blocks of the currentregulator module 170 can be disabled or enabled to change its topologyso that it functions as either the complex vector current regulatormodule 170-1 or the state feedback decoupling current regulator module170-2. In addition, as will also be described below, decoupling voltagesthat are applied are changed based on whether the topology of thecurrent regulator module 170 functions as either a complex vector (CV)current regulator module 170-1 or a state feedback decoupling (SFbD)current regulator module 170-2.

FIG. 4A is a block diagram of a current regulator 170-1, such as thecurrent regulator 170 shown in FIG. 2, configured in accordance with oneembodiment. FIG. 4A will be described with continued reference to FIG.2. The current regulator 170 is a complex vector current regulatorhaving a d-axis regulating portion 302 and a q-axis regulating portion304 with cross-coupling between these portions. The d-axis regulatingportion 302 receives the synchronous reference frame d-axis currentsignal (Id) 132, and the d-axis current command (Id*) 142 and generatesa d-axis current error (Iderror) 311, and the q-axis regulating portion304 receives the synchronous reference frame q-axis current signal (Iq)134 and the q-axis current command (Iq*) 144 and generates a q-axiscurrent error (Iqerror) 331. Each of the regulating portions 302 and 304produces a synchronous frame commanded voltage (e.g., the synchronousreference frame d-axis voltage command signal (Vd*) 172 and asynchronous reference frame q-axis voltage command signal (Vq*) 174) aswill be described below.

Block 320 applies a proportional gain (Kpd) to the d-axis current error(Iderror) 311 to scale the d-axis current error (Iderror) 311 andgenerate a d-axis proportional term 321, which is a scaled value of thed-axis current error (Iderror) 311 scaled by the proportional gain(Kpd). Block 314 applies an integral gain (Kid) to the d-axis currenterror (Iderror) 311 to scale the d-axis current error (Iderror) 311 andgenerate a d-axis integral term 315, which is a scaled value of thed-axis current error (Iderror) 311 scaled by the integral gain (Kid).Block 344 applies a complex gain (ω_(e)Kppd) to the q-axis current error(Iqerror) 331 to scale the q-axis current error (Iqerror) 331 andgenerate an output 345, which is a scaled value of the q-axis currenterror (Iqerror) 331 scaled by the complex gain (ω_(e)Kppd). In oneembodiment, the value of We is the rotor flux speed in electrical rad/s.The summing block 316 combines the d-axis integral term 315 and output345 to generate a d-axis integrator input 317. The integrator 318integrates the d-axis integrator input 317 to generate a d-axisintegration term (Itermd) 319, which is a voltage. The summing block 322combines the d-axis proportional term 321 and the d-axis integrationterm (Itermd) 319 to generate the synchronous reference frame d-axis PIoutput signal 323. The summing block 328 combines the signal 323 and ad-axis decoupling voltage (Vdcpld) 326 to generate the synchronousreference frame d-axis voltage command signal (Vq*) 172.

Block 340 applies a proportional gain (Kpq) to the q-axis current error(Iqerror) 331 to scale the q-axis current error (Iqerror) 331 andgenerate a q-axis proportional term 341, which is a scaled value of theq-axis current error (Iqerror) 331 scaled by the proportional gain(Kpq). Block 334 applies an integral gain (Kiq) to the q-axis currenterror (Iqerror) 331 to scale the q-axis current error (Iqerror) 331 andgenerate a q-axis integral term 335, which is another scaled value ofthe q-axis current error (Iqerror) 331 scaled by the integral gain(Kiq). Block 324 applies a complex gain (ω_(e)Kppq) to the d-axiscurrent error (Iderror) 311 to scale the d-axis current error (Iderror)311 and generate an output 325, which is a scaled value of the d-axiscurrent error (Iderror) 311 scaled by the complex gain (ω_(e)Kppq). Thesumming block 336 combines the q-axis integral term 335 and output 325to generate a q-axis integrator input 337. The integrator 338 integratesthe q-axis integrator input 337 to generate a q-axis integration term(Itermq) 339. The summing block 342 combines the q-axis proportionalterm 341 and the q-axis integration term (Itermq) 339 to generate asynchronous reference frame q-axis PI output signal 343. The summingblock 348 combines the signal 343 and a q-axis decoupling voltage(Vdcplq) 346 to generate the synchronous reference frame q-axis voltagecommand signal (Vq*) 174.

As illustrated in FIG. 4B, the complex gain block 344 includes a gainterm (Kppd) that can be set to zero to disable the complex gain block344 and effectively remove it from the d-axis regulating portion 302 sothat it is no longer part of the d-axis regulating portion 302.Similarly, the complex gain block 324 includes a gain term (Kppq) thatcan be set to zero to disable the complex gain block 324 and effectivelyremove it from the q-axis regulating portion 304 so that it is no longerpart of the q-axis regulating portion 304. When the complex gain blocks324, 344 are removed from the complex vector current regulator module170-1 that is illustrated in FIG. 4A, the complex gain blocks 324, 344have no impact on the current regulation performed by the currentregulator 170, and as illustrated in FIG. 4B, the current regulatormodule 170-2 then functions as a state feedback decoupling currentregulator module 170-2 (as opposed to functioning as a complex vectorcurrent regulator module 170-1 that is illustrated in FIG. 4A).

Depending on which mode the current regulator 170 is operating in at anyparticular time, the d-axis decoupling voltage (Vdcpld) 326, 329 and theq-axis decoupling voltage (Vdcplq) 346, 347 are different. For example,when the current regulator 170 functions as the SFbD current regulatormodule 170-2 (FIG. 4B) for an IPMSM, the d-axis decoupling voltage(Vdcpld) 329 and the q-axis decoupling voltage (Vdcplq) 347 are shown inequations (1A) and (2A) as follows:

Vdcpld=−ω _(e) L _(q) I _(q)   (1A)

Vdcplq=ω _(e)λ_(pm)+ω_(e) L _(d) I _(d)   (2A)

By contrast, when the current regulator 170 functions as the CV currentregulator module 170-1 (FIG. 4A) for an IPMSM, the d-axis decouplingvoltage (Vdcpld) 326 and the q-axis decoupling voltage (Vdcplq) 346 areshown in equations (3A) and (4A) as follows:

Vdcpld=0   (3A)

Vdcplq=ω_(e)λ_(pm)   (4A)

As another example, when the current regulator 170 is operating as theSFbD current regulator module 170-2 (FIG. 4B) for an induction machine,the d-axis decoupling voltage (Vdcpld) 329 and the q-axis decouplingvoltage (Vdcplq) 347 are shown in equations (1B) and (2B) as follows:

$\begin{matrix}{{Vdcpld} = {{{- \omega_{e}}L_{s}\sigma \; I_{q}} - {\omega_{r}\frac{L_{m}}{L_{r}}\lambda_{dr}}}} & \left( {1B} \right) \\{{Vdcplq} = {{{- \omega_{e}}L_{s}\sigma \; I_{d}} + {\frac{R_{r}L_{m}}{L_{r}^{2}}\lambda_{dr}}}} & \left( {2B} \right)\end{matrix}$

and when the current regulator 170 functions as the CV current regulatormodule 170-1 (FIG. 4A) for an induction machine, the d-axis decouplingvoltage (Vdcpld) 326 and the q-axis decoupling voltage (Vdcplq) 346 areshown in equations (3) and (4) as follows:

$\begin{matrix}{{Vdcpld} = {{- \omega_{r}}\frac{L_{m}}{L_{r}}\lambda_{dr}}} & \left( {3B} \right) \\{{Vdcplq} = {\frac{R_{r}L_{m}}{L_{r}^{2}}\lambda_{dr}}} & \left( {4B} \right)\end{matrix}$

In equations (1B), (2B), (3B) and (4B), R_(r) is the rotor resistance,L_(r) is the rotor inductance, L_(m) is the mutual inductance, ω_(r) isthe rotor speed (electrical rad/s), ω_(e) is the rotor flux speed(electrical rad/s), L_(s) σ is the stator transient inductance, andλd_(r) is rotor flux (in field oriented frame).

In addition, at the instant when the current regulator 170 switches fromfunctioning as the SFbD current regulator module 170-2 (FIG. 4B) to theCV current regulator module 170-1 (FIG. 4A), integration terms (Item)applied at the integrators 318, 338 can be re-initialized or changed asshown in (5) and (6) as follows:

Iterm_(d) =Iterm_(d) +V _(dcpldprevious) −V _(dcpld)   (5)

Iterm_(q) =Iterm_(q) +V _(dcplqprevious) −V _(dcplq)   (6)

The integration terms (Iterm) 319, 339 in equations (5) and (6) areapplied at the integrators 318, 338 only at the instant switching occursfrom operating as the SFbD current regulator module 170-2 (FIG. 4B) tothe CV current regulator module 170-1 (FIG. 4A), or vice-versa (e.g., atthe instant switching occurs from operating as the CV current regulatormodule 170-1 (FIG. 4A) to the SFbD current regulator module 170-2 (FIG.4B)).

In equations (1A)-(4A), ω_(e) is the electrical fundamental synchronousfrequency of the machine (in radians per second), I_(d) and I_(q) arethe synchronous reference frame d-axis and q-axis current signals 132,134, λ_(pm) is the permanent magnet flux linkage, respectively, whichare functions of the d-axis and q-axis synchronous frame statorcurrents, L_(d) and L_(q) are the d-axis and q-axis stator inductances,which are functions of the synchronous reference frame d-axis and q-axiscurrent signals 132, 134.

In addition, it should be noted that in some embodiments, but not allembodiments, a virtual damping resistance (that will be described ingreater detail below) can also be used in equations (1) through (4) tocontribute to the d-axis decoupling voltage (Vdcpld) 326 and the q-axisdecoupling voltage (Vdcplq) 346. For example, equations (1A), (1B), (3A)and (3B) could be modified to subtract a correction factor Rdampld, andequations (2A), (2B), (4A) and (4B) could be modified to subtract acorrection factor RdampIq.

Thus, as described above with reference to FIGS. 4A and 4B, the currentregulator can be configured to operate in a first operational mode (asthe CVCR 170-1) when the speed is greater than a first speed threshold(ω₂), or can be configured to operate in a second operational mode (asthe SFbD 170-2) when the speed is below a second speed threshold (ω₁).When the speed is greater than a first speed threshold (ω₂), the currentregulator is configured to operate in the first operational mode as aCVCR 170-1 and the cross-coupling gains (Kppd, Kppq) can be set tonon-zero values. In one embodiment, the values can be set as follows:Kppd=ωb*Lq; Kppq=ωb*Ld. Here, L_(d) and L_(q) are the d-axis and q-axisstator inductances, and ω_(b) represents the commanded bandwidth of thecurrent regulator. When the speed is below the second speed threshold(ω₁), the current regulator is configured to operate in the secondoperational mode as a SFbD 170-2 and the cross-coupling gains (Kppd,Kppq) can be set to zero. This allows a structure of the currentregulator to be modified on-the-fly by toggling certain features. Inaddition to changing the values of the cross-coupling gains (Kppd,Kppq), and the decoupling voltages (Vdcpld, Vdcplq), depending on whichmode the current regulator is operating in, values of the integrationterms (Itermd, Itermq) 319, 339 can be re-initialized of to ensure asmooth voltage output 172, 174 during transitions between operationalmodes. As described above, values of the d-axis and q-axis decouplingvoltages (Vdcpld, Vdcplq) 326, 346 can be varied depending on which modethe current regulator is currently operating in.

FIG. 5 is a flowchart that illustrates a control method 400 that can beapplied to at a current regulator 170 of FIGS. 4A and 4B in accordancewith various embodiments. FIG. 5 will be described with continuedreference to FIGS. 1-4B. The control method 400 can be performed by ahigh-level controller 112 and current regulator 170 of FIG. 2 inaccordance with the present disclosure. Method 400 allow an operationalmode of a current regulator 170 to be configured before executing thecurrent regulator 170 (at 426) in that operational mode. As will beexplained below, the method 400 is used to determine values for thecross-coupling gains (Kppd, Kppq), and the decoupling voltages (Vdcpld,Vdcplq) 326, 346 for the next execution cycle of the current regulator170 before it is executed in accordance with the particular operationalmode that it is currently configured to operate in. Furthermore, method400 will reinitialize the integration terms (Itermd, Itermq) 319, 339 atthe instant that the operational mode of the current regulator 170 ismodified. As can be appreciated in light of the disclosure, the order ofoperation within the method is not limited to the sequential executionas illustrated in FIG. 5, but may be performed in one or more varyingorders as applicable and in accordance with the present disclosure. Invarious embodiments, the method 400 can be scheduled to run based on oneor more predetermined events, and/or can run continuously duringoperation of the vehicle 1.

When the method 400 starts at 402, at 404, previous values of thedecoupling voltages (Vdcpld′, Vdcplq′) 326′, 346′ are stored along witha previous operational mode (CVCR 170-1 or SFbD 170-2) that the currentregulator 170 has been set to.

The method then proceeds to 406, where a determination is made regardingwhether conditions have been satisfied for the current regulator 170 tooperate in a first configuration (as a CVCR 170-1). In one embodiment,speed (e.g., motor speed or electrical fundamental frequency) iscompared to a first speed threshold (ω₂). When the speed is greater thanthe first speed threshold (ω₂), the method 400 proceeds to 408, thecurrent regulator 170 is placed in the first operational mode to operatein a first configuration (as a CVCR 170-1). When the speed is less thanor equal to the first speed threshold (ω₂), the method proceeds to 410.

At 410, a determination is made regarding whether conditions have beensatisfied for the current regulator 170 to operate in a secondconfiguration (as a SFbD 170-2). In one embodiment, speed is compared toa second speed threshold (ω₁). When the speed is less than the secondspeed threshold (ω₁), the method 400 proceeds to 412, where the currentregulator 170 is placed in the second operational mode to operate in thesecond configuration (as a SFbD 170-2).

When the speed is greater than or equal to the second speed threshold(ω₁), the method proceeds to 414, where the current regulator 170remains in its current operational mode to operate in a currentconfiguration (as a CVCR 170-1 or SFbD 170-2) depending on the previousoperational mode (CVCR 170-1 or SFbD 170-2) that the current regulator170 was currently set to at 404.

Following steps for 408, 412, and 414, the method 400 proceeds to 416,where the operational mode of the current regulator 170 is applied. Whenthe operational mode of the current regulator 170 is in the firstoperational mode (as the CVCR 170-1), the method proceeds to 418. At418, the cross-coupling gains (Kppd, Kppq) are enabled and the values ofthe decoupling voltages (Vdcpld, Vdcplq) are updated (as describedabove) from the previous values to configure the current regulator 170in the first operational mode (as the CVCR 170-1).

When the operational mode of the current regulator 170 is in the secondoperational mode (as the SFbD 170-2), the method proceeds to 420. At420, the cross-coupling gains (Kppd, Kppq) are disabled and the valuesof the decoupling voltages (Vdcpld, Vdcplq) 326, 346 are updated (asdescribed above) from the previous values to configure the currentregulator 170 in the second operational mode (as the SFbD 170-2). Thus,when operating as the SFbD current regulator module, the cross-couplinggains (Kppd, Kppq) are set to 0, and when operating as the CV currentregulator module cross-coupling gains (Kppd, Kppq) are tuned accordinglyto a non-zero value as described above.

After 418 and 420, the method 400 proceeds to 422 where a determinationis made whether the operational mode of the current regulator 170 haschanged at 416 (i.e., during this execution of method 400 the previousoperational mode of the current regulator stored in 404 is not equal tothe present operational mode of the current regulator set in 408 or412). When it is determined (at 422) that the operational mode of thecurrent regulator 170 has not changed (at 416), the method proceedsdirectly to 426. When it is determined (at 422) that the operationalmode of the current regulator 170 has changed (at 416) this means that atransition of operational modes has occurred, and the method proceeds to424, where integration terms (Itermd, Itermq) 319, 339 of the currentregulator 170 are reinitialized at 426 (as described above) beforeexecuting a next cycle of the current regulator 170 to allow for asmooth instantaneous transition between the different operating modes.

At 426, the next execution cycle of the current regulator 170 isexecuted in accordance with the particular operational mode that it iscurrently configured to operate in and with values of the cross-couplinggains (Kppd, Kppq), the integration terms (Itermd, Itermq) 319, 339 andthe decoupling voltages (Vdcpld, Vdcplq) 326, 346 set to their presentlydetermined values for that execution cycle.

The execution cycle of the method 400 ends at 428, but it should beappreciated that this method 400 can run continuously to adjust theoperational mode of the current regulator 170 based on speed, and thatFIG. 5 simply shows one iteration of the method 400.

FIG. 6 is a graph that illustrates how the current regulator can beconfigured to operate in a first operational mode (as the CVCR 170-1)when the speed is greater than or equal to a first speed threshold (ω₂),or can be configured to operate in a second operational mode (as theSFbD 170-2) when the speed is less than or equal to a second speedthreshold (ω₁) in accordance with the disclosed embodiments. In oneembodiment, the speed referenced in FIG. 6 is a fundamental electricalfrequency, but in other embodiments speed could also be mechanical shaftspeed. As noted above, this allows a structure of the current regulatorto be modified on-the-fly by toggling certain features. In particular,FIG. 6 shows a hysteresis plot, where above first speed threshold (ω₂),the CVCR will always be used, below the second speed threshold (ω₁) SFbDwill always be used, but between the second speed threshold (ω₁) and thefirst speed threshold (ω₂) the active current regulator will never bechanged. For example, if operating in SFbD, the active current regulatorconfiguration will not change to CVCR until a speed higher than thefirst speed threshold (ω₂) is reached, and the active current regulatorconfiguration will not change back to SFbD until a speed lower than thesecond speed threshold (ω₁) is reached. This means that between thesecond speed threshold (ω₁) and the first speed threshold (ω₂), thecurrently active current regulator configuration will be maintained.

In addition, it is noted that the sampling frequency of the controllerand switching frequency of the inverter can vary considerably duringoperation of an electric machine. It would be desirable to optimize thevalue of the virtual damping resistance that is applied at the currentregulator under all operating conditions as sampling and switchingfrequencies vary during operation. For example, high virtual dampingresistance is generally desired in AC current regulation because it canreduce current regulator parameter sensitivity and can increase currentregulator dynamic stiffness thereby improving overall current regulatorrobustness.

To address this issue, embodiments will now be described that can allowfor the value of virtual damping resistance to vary as a function ofswitching frequency. This allows for the highest possible value ofvirtual damping resistance to be utilized at the current regulator inall switching frequency operating conditions. This approach can bebeneficial, for example, because increasing the value of virtual dampingresistance reduces current regulator parameter sensitivity and increasescurrent regulator dynamic stiffness thereby improving overall currentregulator robustness.

In accordance with some of the disclosed embodiments, a currentregulator is provided for an electric machine drive system for drivingan electric machine. The current regulator includes an adjustabledamping module that has a value of virtual damping resistance that isapplied at the current regulator. The value of virtual dampingresistance is adjustable as a function of sampling frequency. Acontroller can control the current regulator by determining whether thesampling frequency has changed since a previous execution cycle of thecurrent regulator, and when the sampling frequency has changed since theprevious execution cycle, the controller can modify the damping value asa function of the sampling frequency to allow the damping value tochange with the sampling frequency. The damping value has a new value ofvirtual damping resistance that is applied at the current regulatorafter modifying the damping value. The controller can then execute thecurrent regulator in accordance with the modified damping value togenerate the voltage commands. Thus, in these embodiments, a virtualdamping factor that is applied at the current regulator module 170 canbe varied based on or changed as a function of sampling frequency.

FIG. 7 is a block diagram of a current regulator 770 in accordance withanother implementation of the disclosed embodiments. The currentregulator 770 of FIG. 7 includes many of the same blocks as the currentregulators 170-1, 170-2 of FIGS. 4A and 4B, and the descriptions ofFIGS. 4A and 4B are equally applicable to FIG. 7. As such, any elementsof FIGS. 4A and 4B that function the same as corresponding elements ofFIG. 7 are labeled in FIG. 7 with the same reference numbers used inFIGS. 4A and 4B. Any elements of FIG. 7 that perform or operatedifferently than those shown in FIGS. 4A and 4B are labeled in FIG. 7with reference numbers that begin with 7. Thus, although not illustratedin FIG. 4A and 4B, it should be appreciated that in some embodiments,the current regulator 170 can be also include features of FIG. 7 so thatit also implements a virtual damping resistance that can be modified asa function of sampling frequency (F_(S)). The sampling frequency (F_(s))refers to the rate at which the controller samples feedback signals,performs control calculations, and updates its outputs. The samplingfrequency (F_(s)) is expressed in hertz (Hz). Depending on theimplementation, this sampling frequency (F_(s)) can be proportional orequal to switching frequency of the inverter module. In this regard itis noted that the sampling frequency (F_(s)) can be the same or amultiple of the switching frequency (F_(SW)) so that synchronizationbetween sampling and PWM can be achieved.

In this embodiment, damping value (Rdamp) 710 is multiplied by thesynchronous reference frame d-axis current signal (Id) 132 and to thesynchronous reference frame q-axis current signal (Iq) 134 to generatedamped current signals 712, 722. In accordance with the disclosedembodiments, the controller 112 (of FIG. 2) can vary the damping value(Rdamp) 710 based on the sampling frequency (F_(s)) whenever thesampling frequency changes. The damping value (Rdamp) 710 can be updatedregularly based on sampling frequency (F_(s)).

The integrator 318 integrates the d-axis integrator input 317 togenerate a d-axis integration term (Itermd) 719, and the integrator 338integrates the q-axis integrator input 337 to generate a q-axisintegration term (Itermq) 739. As the damping value transitions, thecontroller 112 can also reinitialize integration terms (Itermd, Itermq)719, 739 before the current regulator 770 executes (for the presentexecution cycle) to help ensure smooth synchronous reference framevoltage command signals (Vd*, Vq*) 172, 174 while the damping value isbeing updated to the new damping value (Rdamp). Notably, applying thegains 314, 324, 334, 344 before integration process at 318, 338 allowsfor easier reinitialization of integration terms (Itermd, Itermq) 719,739 so that the transition to the new damping value (Rdamp) goessmoothly and does not induce disturbances into the synchronous referenceframe voltage command signals (Vd*, Vq*) 172, 174 during the transition.

In one embodiment, to reinitialize the integration terms (Itermd,Itermq) 719, 739, the controller 112 stores the previous damping valuefor use in further computations when it updates the new damping value(Rdamp) based on the newest sampling frequency (F_(s)). For example, inone embodiment, a change in the damping value (ΔRdamp) is calculated bysubtracting the previous damping value (Rdamp′) from the new dampingvalue (Rdamp), and while the previous damping value (Rdamp′) is beingupdated to the new damping value (Rdamp), the integration terms (Itermd,Itermq) 719, 739 of the current regulator 770 are reinitialized toensure a smooth voltage output. The integration terms (Itermd, Itermq)719, 739 of the current regulator 770 can be reinitialized in accordancewith equations (7) and (8) as follows:

Iterm_(d) =Iterm_(d,previous+ΔR)damp I _(d)   (7)

Iterm_(q) =Iterm_(q,previous) +ΔRdamp I _(q)   (8)

In this embodiment, the summing block 328 combines the signal 323 andthe damping voltage signal 712 to generate the synchronous referenceframe d-axis voltage command signal (Vq*) 172. In one embodiment, thed-axis decoupling voltage (Vdcpld) (not shown in FIG. 7) is set to zero.The summing block 348 combines the signal 343, the damping voltagesignal 722, and a q-axis decoupling voltage (Vdcplq) 724 to generate thesynchronous reference frame q-axis voltage command signal (Vq*) 174. Inone embodiment, the q-axis decoupling voltage (Vdcplq) 724 is set to theback-EMF voltage (e.g., ωeλ_(pm) for a synchronous machine).

FIG. 8 is a flowchart that illustrates a control method 800 that can beapplied to at a current regulator 770 of FIG. 7 in accordance withvarious embodiments. In particular, FIG. 8 illustrates a method 800 formodifying a damping value (Rdamp) 710 of a current regulator 770 as afunction of, for example, sampling frequency (F_(s)) and then executingthe current regulator 770 in accordance with the disclosed embodiments.However, it should be appreciated that other equivalent measures ofsampling frequency (F_(s)) could be utilized in conjunction with themethod 800 including those described above. As will be explained below,the method 800 is used to set the damping value (Rdamp) 710 as afunction of sampling frequency, and to set appropriate integration terms(Itermd, Itermq) 719, 739 for the next execution cycle of the currentregulator 770 before it is executed to ensure a smooth output voltage172, 174 during a transition period.

When the method 800 starts at 802, at 804, it is determined whether thesampling frequency (F_(s)) has changed. When it is determined (at 804)that the sampling frequency (F_(s)) has not changed, the method 800proceeds to 814, where the current regulator 770 is executed.

When it is determined (at 804) that the sampling frequency (F_(s)) haschanged, the method 800 proceeds to 806. Steps 806 through 812 areperformed to update the damping value (Rdamp) 710 based on samplingfrequency (F_(s)) and to reinitialize integration terms (Itermd, Itermq)719, 739 before the current regulator 770 is executed at 814 to ensure asmooth voltage output after the damping value has been updated to thenew damping value (Rdamp) that was calculated at 808.

At 806, the previous damping value is stored for use in furthercomputations. At 808, the new damping value (Rdamp) is updated based onthe present sampling frequency (F_(s)). At 810, a change in the dampingvalue is calculated by subtracting the previous damping value (Rdamp′)from the new damping value (Rdamp).

At 812, the integration terms (Itermd, Itermq) 719, 739 of the currentregulator 770 are reinitialized to ensure no disturbance at the outputvoltage occurs when the new damping value (Rdamp) that was calculated at808 is applied.

At 814, the next execution cycle of the current regulator 770 isexecuted in accordance with the updated value of the damping value(Rdamp) 710 and updated values of the integration terms (Itermd, Itermq)719, 739 set to their presently determined values for that executioncycle. The execution cycle of the method 800 ends at 816, but it shouldbe appreciated that this method 800 runs continuously to adjust thecurrent regulator and that FIG. 8 simply shows one iteration of themethod 800.

FIG. 9 is graph that illustrates an example of how virtual dampingresistance (Rdamp) could vary as a function sampling frequency (F_(S))in accordance with the embodiment illustrated in FIGS. 7 and 8. As thesampling frequency (F_(S)) increases, the virtual damping resistance(Rdamp) increases, and conversely, as the sampling frequency (F_(S))decreases, the virtual damping resistance (Rdamp) decreases.

Thus, various embodiments have been described for current regulatorsthat can be used for controlling operation of a multi-phase machine in avector controlled motor drive system. The disclosed embodiments providea current regulator 170 that can be configured to operate in a firstoperational mode (as the CVCR 170-1) when the speed (e.g., motor speedor electrical fundamental frequency) is greater than a first speedthreshold (ω₂), or can be configured to operate in a second operationalmode (as the SFbD 170-2) when the speed is greater than a second speedthreshold (ω₁). This allows a structure of the current regulator to bemodified on-the-fly by toggling certain features. The disclosedembodiments can also provide a mechanism for modifying a damping value(Rdamp) of a current regulator as a function of sampling frequency(F_(s)).

In other embodiments, a current regulator is provided that allows forthe value of virtual damping resistance to vary as a function ofswitching frequency. This allows for the highest possible value ofvirtual damping resistance to be utilized at the current regulator inall switching frequency operating conditions. This current regulator canhelp optimize the value of the virtual damping resistance that isapplied at the current regulator under all operating conditions assampling and switching frequencies vary during operation. This currentregulator can reduce parameter sensitivity and can increase dynamicstiffness thereby improving overall robustness of the current regulator.

Those of skill in the art would further appreciate that the variousillustrative logical blocks, modules, circuits, and algorithm stepsdescribed in connection with the embodiments disclosed herein may beimplemented as electronic hardware, computer software, or combinationsof both. Some of the embodiments and implementations are described abovein terms of functional and/or logical block components (or modules) andvarious processing steps. However, it should be appreciated that suchblock components (or modules) may be realized by any number of hardware,software, and/or firmware components configured to perform the specifiedfunctions.

To clearly illustrate this interchangeability of hardware and software,various illustrative components, blocks, modules, circuits, and stepshave been described above generally in terms of their functionality.Whether such functionality is implemented as hardware or softwaredepends upon the particular application and design constraints imposedon the overall system. Skilled artisans may implement the describedfunctionality in varying ways for each particular application, but suchimplementation decisions should not be interpreted as causing adeparture from the scope of the present invention. For example, anembodiment of a system or a component may employ various integratedcircuit components, e.g., memory elements, digital signal processingelements, logic elements, look-up tables, or the like, which may carryout a variety of functions under the control of one or moremicroprocessors or other control devices. In addition, those skilled inthe art will appreciate that embodiments described herein are merelyexemplary implementations.

The various illustrative logical blocks, modules, and circuits describedin connection with the embodiments disclosed herein may be implementedor performed with a general purpose processor, a digital signalprocessor (DSP), an application specific integrated circuit (ASIC), afield programmable gate array (FPGA) or other programmable logic device,discrete gate or transistor logic, discrete hardware components, or anycombination thereof designed to perform the functions described herein.A general-purpose processor may be a microprocessor, but in thealternative, the processor may be any conventional processor,controller, microcontroller, or state machine. A processor may also beimplemented as a combination of computing devices, e.g., a combinationof a DSP and a microprocessor, a plurality of microprocessors, one ormore microprocessors in conjunction with a DSP core, or any other suchconfiguration.

The steps of a method or algorithm described in connection with theembodiments disclosed herein may be embodied directly in hardware, in asoftware module executed by a processor, or in a combination of the two.A software module may reside in RAM memory, flash memory, ROM memory,EPROM memory, EEPROM memory, registers, hard disk, a removable disk, aCD-ROM, or any other form of storage medium known in the art. Anexemplary storage medium is coupled to the processor such the processorcan read information from, and write information to, the storage medium.In the alternative, the storage medium may be integral to the processor.The processor and the storage medium may reside in an ASIC. The ASIC mayreside in a user terminal. In the alternative, the processor and thestorage medium may reside as discrete components in a user terminal.

In this document, relational terms such as first and second, and thelike may be used solely to distinguish one entity or action from anotherentity or action without necessarily requiring or implying any actualsuch relationship or order between such entities or actions. Numericalordinals such as “first,” “second,” “third,” etc. simply denotedifferent singles of a plurality and do not imply any order or sequenceunless specifically defined by the claim language. The sequence of thetext in any of the claims does not imply that process steps must beperformed in a temporal or logical order according to such sequenceunless it is specifically defined by the language of the claim. Theprocess steps may be interchanged in any order without departing fromthe scope of the invention as long as such an interchange does notcontradict the claim language and is not logically nonsensical.

Furthermore, depending on the context, words such as “connect” or“coupled to” used in describing a relationship between differentelements do not imply that a direct physical connection must be madebetween these elements. For example, two elements may be connected toeach other physically, electronically, logically, or in any othermanner, through one or more additional elements.

While at least one exemplary embodiment has been presented in theforegoing detailed description, it should be appreciated that a vastnumber of variations exist. It should also be appreciated that theexemplary embodiment or exemplary embodiments are only examples, and arenot intended to limit the scope, applicability, or configuration of thedisclosure in any way. Rather, the foregoing detailed description willprovide those skilled in the art with a convenient road map forimplementing the exemplary embodiment or exemplary embodiments. Itshould be understood that various changes can be made in the functionand arrangement of elements without departing from the scope of thedisclosure as set forth in the appended claims and the legal equivalentsthereof.

1. A method for controlling a current regulator of an electric machinedrive system for driving an electric machine, the method comprising:determining, at a controller, whether a sampling frequency has changedsince a previous execution cycle of the current regulator; storing aprevious damping value that is a previous value of the virtual dampingresistance applied during the previous execution cycle of the currentregulator; modifying, at the controller when the sampling frequency haschanged since the previous execution cycle, a damping value of thecurrent regulator as a function of sampling frequency to allow thedamping value to change with the sampling frequency, wherein the dampingvalue has a new value of virtual damping resistance that is applied atthe current regulator after modifying, wherein the modifying comprises:determining the modified damping value based on the sampling frequencyduring a current execution cycle of the current regulator, wherein themodified damping value is a new value of virtual damping resistancebased on a new sample frequency; computing a change in damping valuebased on the difference between the modified damping value and theprevious damping value, wherein the change in the damping value is achange in the virtual damping resistance based on a difference betweenthe new value of virtual damping resistance and the previous value ofvirtual damping resistance; and executing the current regulator togenerate voltage commands in accordance with the modified damping value.2. The method according to claim 1, wherein the modifying comprises:setting the damping value as a function of the sampling frequency. 3.The method according to claim 1, wherein the modifying comprises:updating the damping value based on the sampling frequency.
 4. Themethod according to claim 1, wherein the modifying comprises: computingthe new value of virtual damping resistance using an equation.
 5. Themethod according to claim 1, wherein the modifying comprises:determining the new value of virtual damping resistance via a lookuptable.
 6. (canceled)
 7. The method according to claim 1, furthercomprising: prior to executing the current regulator, reinitializingintegration terms generated by integrators of the current regulator whenthe virtual damping resistance is updated.
 8. The method according toclaim 1, further comprising: prior to executing the current regulatorduring a current execution cycle of the current regulator,reinitializing integration terms generated by integrators of the currentregulator during a transition period to avoid inducing disturbancevoltages as when a previous damping value is being updated to themodified damping value.
 9. The method according to claim 7, whereinexecuting the current regulator in accordance with the modified dampingvalue, comprises: executing the current regulator using the modifieddamping value and updated values of the integration terms generated byintegrators of the current regulator by setting each to presentlydetermined values to generate voltage commands for the current executioncycle of the current regulator.
 10. The method according to claim 1,further comprising: generating current error values, wherein eachcurrent error value is determined based on a difference between acurrent command value and a stator current value from the electricmachine; applying, at the current regulator, gains to the current errorvalues prior to providing the current error values to integrators of thecurrent regulator that generate integration terms.
 11. The methodaccording to claim 1, wherein the electric machine comprises machineterminals, the method further comprising: generating stationaryreference frame voltage command values by performing a dq-to-αβtransformation on the voltage commands; generating phase voltage commandvalues based on the stationary reference frame voltage command values;generating switching vector signals based on the phase voltage commandvalues; generating three-phase alternating current voltage signalwaveforms based on switching vector signals and a DC input voltage; andapplying the three-phase alternating current voltage signal waveforms tothe machine terminals.
 12. An electric machine drive system for drivingan electric machine, comprising: a current regulator configured togenerate voltage command values, the current regulator comprising: anadjustable damping module that has a value of virtual damping resistancethat is applied at the current regulator, wherein the value of virtualdamping resistance is adjustable as a function of sampling frequency;and a controller that is configured to control the current regulator,the controller being configured to: determine whether the samplingfrequency has changed since a previous execution cycle of the currentregulator; store a previous damping value that is a previous value ofthe virtual damping resistance applied during the previous executioncycle of the current regulator; modify, when the sampling frequency haschanged since the previous execution cycle, the damping value as afunction of the sampling frequency to allow the damping value to changewith the sampling frequency, wherein the damping value has a new valueof virtual damping resistance that is applied at the current regulatorafter modifying the damping value, wherein the modified damping value isdetermined based on the sampling frequency during a current executioncycle of the current regulator, wherein the modified damping value is anew value of virtual damping resistance based on a new sample frequency;compute a change in damping value based on the difference between themodified damping value and the previous damping value, wherein thechange in the damping value is a change in the virtual dampingresistance based on a difference between the new value of virtualdamping resistance and the previous value of virtual damping resistanceand execute the current regulator to generate the voltage commands inaccordance with the modified damping value.
 13. (canceled)
 14. Theelectric machine drive system according to claim 12, wherein the currentregulator further comprises: integrators; and wherein the controller isfurther configured to: reinitialize, prior to executing the currentregulator, integration terms generated by the integrators of the currentregulator based on the change in the virtual damping resistance.
 15. Theelectric machine drive system according to claim 14, wherein the currentregulator is further configured to: generate voltage commands for thecurrent execution cycle based on the modified damping value and updatedvalues of integration terms generated by the integrators by setting eachto presently determined values.
 16. The electric machine drive systemaccording to claim 15, wherein the current regulator is furtherconfigured to: generate current error values, wherein each current errorvalue is determined based on a difference between a current commandvalue and a stator current value from the electric machine; apply gainsto the current error values prior to providing the current error valuesto the integrators that generate integration terms.
 17. The electricmachine drive system according to claim 12, further comprising: asynchronous-to-stationary transformation module that generatesstationary reference frame voltage command values by performing adq-to-αβ transformation on the voltage command values; an αβ-to-abctransformation module that receives the stationary reference framevoltage command values, and generates phase voltage command values; apulse width modulation module that generates switching vector signalsbased on the phase voltage command values; an inverter module thatgenerates three-phase alternating current voltage signal waveforms basedon switching vector signals and a DC input voltage; and an electricmachine comprising machine terminals, wherein the electric machine iscoupled to the inverter module and the three-phase alternating currentvoltage signal waveforms are applied to the machine terminals.
 18. Avehicle comprising a multi-phase electric machine having machineterminals and an electric machine drive system, comprising: a currentregulator configured to generate voltage command values to control themulti-phase electric machine, the current regulator comprising: anadjustable damping module that has a value of virtual damping resistancethat is applied at the current regulator, wherein the value of virtualdamping resistance is adjustable as a function of sampling frequency;and a controller that is configured to control the current regulator,the controller being configured to: determine whether the samplingfrequency has changed since a previous execution cycle of the currentregulator; store a previous damping value that is a previous value ofthe virtual damping resistance applied during the previous executioncycle of the current regulator; modify, when the sampling frequency haschanged since the previous execution cycle, the damping value as afunction of the sampling frequency to allow the damping value to changewith the sampling frequency, wherein the damping value has a new valueof virtual damping resistance that is applied at the current regulatorafter modifying the damping value, wherein the modified damping value isdetermined based on the sampling frequency during a current executioncycle of the current regulator, wherein the modified damping value is anew value of virtual damping resistance based on a new sample frequency;compute a change in damping value based on the difference between themodified damping value and the previous damping value, wherein thechange in the damping value is a change in the virtual dampingresistance based on a difference between the new value of virtualdamping resistance and the previous value of virtual damping resistance;and execute the current regulator to generate the voltage commands inaccordance with the modified damping value.
 19. (canceled)
 20. Thevehicle according to claim 18, wherein the current regulator furthercomprises: integrators; and wherein the controller is further configuredto: reinitialize, prior to executing the current regulator, integrationterms generated by the integrators of the current regulator based on thechange in the virtual damping resistance, and wherein the currentregulator is further configured to: generate current error values,wherein each current error value is determined based on a differencebetween a current command value and a stator current value from theelectric machine; apply gains to the current error values prior toproviding the current error values to the integrators that generateintegration terms.